Bridge-stabilized oscillator circuit and method

ABSTRACT

An apparatus and method are disclosed for improving the stability of the frequency of vibration of an oscillator signal produced by an oscillator circuit. In a preferred embodiment of the present invention, a quartz crystal resonator is one arm of a bridge which generates a bridge signal which varies in accordance with the vibrating frequency of the resonator. A synchronous demodulator responds to the bridge signal for producing an error signal which is converted into a control signal. A control circuit receives the control signal and changes its reactance when the resonator is no longer vibrating at its unperturbed resonance frequency so that the vibration frequency of the resonator connected to the control circuit is returned to its resonant frequency. An automatic level control circuit is also included for controlling the drive level of the signal exciting the resonator.

FIELD OF THE INVENTION

This invention relates to oscillator circuits and, more particularly, tocircuits which incorporate a crystal resonator for generating anoscillator signal having a relatively high level of frequency stability.

BACKGROUND OF THE INVENTION

High precision oscillators are required in many electronic devices as amaster clock or frequency source, from which all other time intervalsand operational frequencies are derived. Quartz crystal resonators areoften used in such oscillators because the resonant frequency of thecrystal is very stable with respect to temperature. Ideally, thefrequency of the oscillator would be the same as the unperturbedresonant frequency of the crystal resonator. Oscillator circuits,however, never oscillate exactly at the unperturbed frequency of thecrystal. The crystal resonator can be fabricated with a high qualityfactor, which is denoted in the art as its Q, so that the resonantfrequency of the crystal is not easily perturbed by associatedcircuitry.

In a prior art component assembly known as the 10811 Oscillator used invarious commercial products manufactured, sold and shipped by theHewlett-Packard Company (HP), the assignee of the present patentapplication, a mode C, SC cut quartz crystal is put into an oven andmaintained at a constant temperature of about 82° C. regardless ofambient temperatures ranging from -30° C. to +70° C. This arrangement,along with the extremely low temperature coefficient of frequency of thecrystal used in the 10811 Oscillator, results in the holding of theunperturbed crystal frequency within about one part in 10¹⁰ over thefull ambient temperature range. The associated circuitry in the 10811Oscillator typically adds frequency perturbations on the order ofseveral parts in 10⁹ over the same ambient temperature range.

However, as modern applications require higher levels of stability, itis relatively difficult to design associated circuitry which maintains asufficiently low degree of perturbation to achieve the requiredstability. Accordingly, there is a need for improving the associatedcircuitry used in oscillators.

SUMMARY OF THE INVENTION

The above-mentioned problems arising in prior art oscillator circuitsare overcome by provision of an oscillator made in accordance with theteachings of the present invention. In a preferred embodiment of thepresent invention, a conventionally known quartz crystal resonator formsone arm of a bridge circuit. The bridge circuit is connected via one ofits ports to a voltage controlled oscillator (VCXO) stage. At an outputof a second port of the bridge circuit is a Radio Frequency (RF) signalwhich is indicative of how close the crystal resonator is to itsresonance level. A demodulator circuit mixes the signals from the twoports of the bridge circuit to produce a direct current (DC) signalwhich is sent to the VCXO stage. If the crystal resonator is vibratingat its resonant frequency, the oscillator continues its operationunchanged. If the resonator no longer vibrates at its resonantfrequency, the voltage controlled oscillator stage responds to theassociated error signal from the demodulator circuit and the reactanceof the voltage controlled oscillator stage is changed so that thevibrating frequency of the resonator is subsequently changed andreturned to its resonant frequency. This arrangement results in anAutomatic Frequency Control (AFC) loop for maintaining stability of theoscillator signal. An Automatic Level Control (ALC) circuit is connectedto the bridge circuit and VCXO stage so that the oscillator operateswith stable amplitude, since non-linearities in the quartz crystal makeits frequency sensitive to the drive power of the signal exciting theresonator.

BRIEF DESCRIPTION OF THE DRAWINGS

The features and advantages of the present invention will be apparent byreference to the following detailed description of the preferredembodiments when taken in conjunction with the following list ofdrawings, where like reference numerals refer to like elements:

FIG. 1A is a schematic of a preferred embodiment of the presentinvention;

FIG. 1B is a schematic of a conventional bipolar transistor RF amplifierused in the oscillator shown in FIG. 1A;

FIG. 2 is a schematic of an expanded embodiment of the present inventionwhich includes a second synchronous demodulator added to the demodulatordepicted in FIG. 1A;

FIG. 3 is a schematic of still another embodiment of the presentinvention which includes an Automatic Level Control (ALC) circuit;

FIG. 4 is an alternate embodiment of the present invention whichincludes a high precision ALC detector and wide range AutomaticFrequency Control (AFC) detector;

FIG. 5. depicts a schematic of an electronic bridge circuit which is analternative to the bridge circuit shown in FIG. 1;

FIG. 6 shows a schematic of a precision phase inverter used in theelectronic bridge circuit depicted in FIG. 5;

FIGS. 7-11 depict schematics of various portions of a working oscillatorcircuit made in accordance with the teachings of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIGS. 1A & 1B depict a preferred embodiment of an oscillator 8 madeaccording to the teachings of the present invention. A conventional RFamplifier 10 is an inverting type with relatively high input and outputimpedances. The output current of the amplifier 10 is sent to aconventional pi-network 11 comprising capacitors 12 and 14 and inductor16 via the primary impedance of a transformer 18. On the other side ofthe transformer 18 is a terminal 20 where an oscillator output signal 22would appear and to which a load impedance 24 is connected. If node 25were grounded and the pi-network 11 were operated at its resonantfrequency, the phase shift of a signal flowing through the network wouldbe 180 degrees. Together with the 180 degree phase shift produced by theamplifier 10, this arrangement results in 0 degree net loop phase shift,which is one of the required conditions for the network to oscillate.Another condition required to initiate oscillation is to have greaterthan unity net loop gain. When the net loop gain is equal to unity, theoscillation is maintained in steady state. This other condition dependson the relationship between the transconductance of an active componentin the amplifier 10, the Q of the inductor 16, and the ratio of thecapacitances of capacitors 12 and 14. As will be described in a laterportion of this description in connection with FIG. 1B, the activeelement of the amplifier 10 is a transistor.

At node 25 is connected one end of a varactor diode 26 (functioning as acapacitor) which is in series with an inductor 28 and together form aseries resonant LC circuit 29. The inductance of the inductor 28 ischosen so that the LC circuit 29 resonates at the resonant frequency ofa quartz crystal resonator 30 when the bias voltage of the varactordiode 26 is in the center of its range. A bridge circuit 31 is formed bythe crystal resonator 30 and an image resistor 32 connected to a hybridtransformer 34. As is the convention, the phase of each coil of thetransformer 34 is indicated by the dots 37. The value of the imageresistor 32 is equal to the Equivalent Series Resistance (ESR) of theresonator 30. An output port 50 of the bridge circuit 31 is connected inseries between the inductor 28 and a ground.

The bias circuit for the varactor diode 26 includes the inductor 28,resistor 32, transformer 34, and a resistor 36, with the capacitors 12and 14 connected for blocking the bias voltage. When the bias voltage isnot in the center of its range, the reactance of the series resonant LCcircuit 29 is either net inductive or net capacitive, which will resultin a pulling of the frequency of oscillation below or above thefrequency of crystal resonance, respectively. In the preferredembodiments of the present invention, the resonator 30 is a quartzcrystal. A more detailed description of such quartz crystal is providedin an article entitled, "SC-Cut Quartz Oscillator Offers ImprovedPerformance," Hewlett-Packard Journal, March 1981, pp. 20-29.

Part of a frequency stabilization loop 35 comprises the resistor 36, LCcircuit 29, an RF amplifier 38, and a mixer 42. The amplifier 38amplifies the signal at the output port 50. Another part of thestabilization loop 35 includes three cascaded tuned RF amplifiers 39, 40and 41 and the mixer 42. The three RF amplifiers amplify the signal froma port 33 of the bridge circuit 31. The output of the mixer 42 isconnected to a resistor 44 which in turn is connected to the invertinginput of an operational amplifier 46 and a capacitor 48. Thenon-inverting input 49 of the operational amplifier 46 is grounded.

The frequency stabilization circuit 35 begins with the secondary windingof the transformer 34. The magnitude and phase of an RF signal 52 whichpasses through the port 33 into the frequency stabilization circuit 35is proportional to the magnitude and phase of the reactance of thecrystal resonator 30. The phase of the signal 52 at the port 33 withrespect to that for a signal 53 at port 50 is -90 (negative ninety)degrees when the net reactance of the crystal resonator 30 is inductive,which occurs when the frequency of oscillation is above the resonantfrequency of the crystal resonator. Similarly, the phase of the signal52 at the port 33 is +90 (positive ninety) degrees when the frequency ofoscillation is below the resonant frequency of the crystal resonator. Inthe very narrow frequency range where the amplitude of the signal 52 isnear its minimum, the phase transitions continuously in a region from+90 to -90 degrees as the frequency is increased. This transition regionis typically a few milliHertz wide. Tuned RF amplifiers 39, 40, and 41amplify the signal 52, which is an error signal from the bridge circuit31. The amplifier 38 amplifies the signal 53, which is a referencesignal from the bridge circuit.

The amplifier 38 is an inverting type so that the phase of a signal 54appearing at one input of the mixer 42 is 180 degrees with respect tothat for the signal 52. Since the amplifiers 39, 40, and 41 are alsobasically inverting amplifiers, the phase of a signal 43 appearing atanother input of the mixer 42 would also be 180 degrees with respect tothe phase of the signal 52 in the absence of any additional phaseshifts. However, the amplifiers 39, 40, and 41 are actually tuned to afrequency preferably a few percent below the crystal frequency such thatan additional 30 degrees phase lag occurs in each amplifier, for a totalof -90 degrees of extra phase shift. Thus, the phase of the signal 43 is-270 degrees (or equivalently +90 degrees) with respect to the signal52.

The result of all of the above described phase shifting is that thephase of the signal 43 with respect to the signal 54 is 180 degrees whenthe frequency of oscillation is above the resonant frequency of thecrystal resonator 30, and 0 degrees when the oscillation frequency isbelow the resonant frequency. Accordingly, a positive DC output signal55 is produced at an inverting output of the mixer 42 when theoscillation frequency is above the resonant frequency and a negative DCoutput signal 55 when the oscillation frequency is below the resonantfrequency. The signal 55 drives the output of an inverting integrator 56in a negative direction which reduces the oscillation frequency when itis above the resonant frequency. In like manner, the signal 55 drivesthe output of the inverting integrator 56 in a positive direction whichincreases the oscillation frequency when it is below the resonantfrequency. In both cases, a so-called servo loop is created whichoperates to drive the oscillation frequency so that it converges to theresonant frequency.

A power supply (not shown) connected to amplifier 10 provides theelectrical energy needed for producing a drive signal which passesthrough the LC circuit 29 for exciting the crystal resonator 30 tovibrate. As long as the input impedance of the amplifier 38 is high, thebridge circuit 31 is essentially transparent at node 50 to theoscillator 8.

The demodulator function of the mixer 42 is insensitive to any errorsignals caused by a mismatch between the image resistor 32 and the ESRof the crystal resonator 30 because the phase of the extraneous signaldue to the mismatch appearing at the RF input 61 of the mixer 42 wouldbe in quadrature with the phase of the signal appearing at the LO input63 of the mixer 42. The effect of drift in the phase shift caused by theamplifiers 39, 40, and 41 is to reduce immunity to resistance imbalance,but in the absence of such imbalance, the only deleterious effect is toreduce the error loop gain by a factor equal to the cosine of the phaseerror. Accordingly, the accuracy of the circuit is not critical withrespect to the phase stability of the amplifiers 38, 39, 40, and 41. AnIF (Intermediate Frequency) signal appearing at an output terminal 65 ofthe mixer 42 is integrated by the integrator 56 to provide a DC controlvoltage that is fed back to the bias circuit of the varactor diode 26via resistor 36 and correspondingly changes the capacitance of thevaractor diode 26. The oscillator 8 is thus kept in tune to thefrequency that balances the bridge circuit 31, which occurs when thecrystal resonator is vibrating at its exact series resonant frequency.

For frequencies more than a few kilohertz away from the resonantfrequency of the resonator 30, the impedance of the bridge circuit 31between node 50 and ground becomes many hundreds of ohms. Since theamplifier 10 preferably includes a grounded emitter bipolar transistor(not shown in FIG. 1A but depicted in FIG. 1B), there is no lowimpedance path from the base (the input of the amplifier 10) to theemitter, which is grounded. Since the two capacitors 12 and 14 onlyconnect to ground through the bridge circuit 31, the high impedance ofthis path prevents any RF base current from flowing so there is nobase-recombination current noise at those frequencies. By inserting theprimary winding of the transformer 18 in series with the collector ofthe amplifier 10, the oscillator signal 22 at the terminal 20 isextracted without introducing a path to ground that would causebase-recombination noise. With base-recombination noise currenteliminated, the remaining noise mechanism in the transistor arises fromcollector shot noise current. This is typically 10 to 15 dB less thanthe base-recombination noise current. Thus, the so called phase noisefloor, being the phase noise at offsets from the carrier of more than afew kHz, is reduced by a like amount. This same principle is used in theaforementioned 10811 Oscillator, except that in the 10811 Oscillator anextra grounded-base buffer amplifier must be added to implement it. Thetechnique taught here eliminates the need for this extra amplifier. Theextra load impedance added in series with the collector by thetransformer 18 is virtually transparent to the operation of theoscillator 8 because of the typically high collector impedance of thetransistor.

Depicted in FIG. 1B is a detailed schematic of a preferred embodiment ofthe amplifier 10 where a capacitor 80 is connected to a resistor 82 andthe base of a conventional bipolar transistor 84. The resistor 82 isconnected to a voltage source V_(BB) (not shown in detail). Thetransistor 84 is connected to a resistor 86 and a capacitor 88 which areboth grounded and result in the grounding of the emitter of thetransistor 84. The transistor 84 is also connected to an inductor 90 anda capacitor 92. The inductor 90 is connected to a voltage source Vcc(not shown in detail).

With reference to FIG. 2, an oscillator 200 is depicted which is analternate preferred embodiment the oscillator 8 shown in FIGS. 1A & 1B.The oscillator 200 includes a second synchronous demodulator which is amixer 202 operating with its RF input signal shifted 90 degrees withrespect to the corresponding RF signal applied to the mixer 42. Thisarrangement is used to compensate for errors in setting the resistor 32to the exact value of the ESR of the crystal resonator 30. In thisembodiment, the resistor 32 is deliberately made slightly smaller thanthe value which would balance the bridge circuit. A voltage controlledresistance 216 is inserted in series with the resistor 32. Hence, thecontrol voltage for the voltage controlled resistance 216 can be used asa fine adjustment. The mixer 202 generates a DC error signalproportional to the difference between the resistance of the imageresistor 32 and the ESR of the crystal resonator 30. It is insensitiveto the reactance of the resonator 30. This is in contrast to the mixer42 which generates an error signal proportional to the reactance of theresonator 30 but is insensitive to resistance. If the mixer 202 detectsa resistive imbalance in the bridge circuit 31, an integrator 215changes the voltage controlled resistance 216 in the correct directionso that the sum of the voltage controlled resistance 216 and theresistor 32 is equal to the ESR of the resonator 30 and the unbalancedcondition is thus eliminated.

In addition to the elements used for the oscillator 8, the oscillator200 includes a conventional 90 degree phase shifter 204 connected to theoutput of the amplifier 41. The phase shifter 204 is preferablyimplemented as two grounded capacitors 203 and 205 and an inductor 207.The output of the 90 degree phase shifter 204 is connected to an inputterminal 206 which is the RF input for the mixer 202. The IF outputterminal 208 of the mixer 202 is connected to the integrator 215 whichis a resistor 210 connected to an inverting input of an operationalamplifier 212 and a capacitor 214. A non-inverting input to theoperational amplifier 212 is grounded and the output is applied to thevoltage controlled resistance 216, which is well known in the prior artin various forms such as PIN diodes and triode region JFETs.

An oscillator 300 is depicted in FIG. 3 which is another preferredembodiment of the present invention. The oscillator 300 is similar tothe oscillator 8 shown in FIG. 1 but includes additional elementsforming an ALC (Automatic Level Control) for controlling the drive levelof the signal exciting the crystal resonator 30. It is known that theresonant frequency of the crystal is sensitive to the drive level beingapplied because of the non-linear intrinsic properties of the quartzmaterial used in the crystal resonator 30. A description of theoperation of the ALC will first be given before a detailed descriptionis provided for its specific components. The ALC operates so that an RFvoltage at node 50 is rectified by a diode detector 305 which produces aDC voltage at a node 307 that is roughly proportional to the peak RFvoltage at the node 50, except for an offset (preferably of about 0.2volts) due to the turn-on voltage of a Schottky diode 302. An integrator309 integrates the DC voltage at the node 307 and applies it to acapacitive voltage divider 311. The voltage at a node 313 determines thedivide ratio of the voltage divider 311 which in turn controls thefeedback around the amplifier 10. By regulating this feedback, theamplitude is controlled so as to servo the RF voltage at the node 50 toa constant value. When the bridge circuit 31 is in balance, thisarrangement will also servo the drive power being applied to the crystal30 to a constant value.

The oscillator 300 further includes a diode 302 having its anodeconnected to the node 50. The cathode of the diode 302 is connected to abypass capacitor 304 and a DC load resistor 306. The cathode of thediode 302 is also connected to a resistor 308 which in turn is connectedto the inverting input of an operational amplifier 310 and a capacitor312. The anode of the diode 302 is connected to ground via the resistor32 and the transformer 34. The non-inverting input of the amplifier 310is connected to a voltage source V_(REF) and the output is connected toa resistance 314 for biasing two varactor diodes 316 and 318. Thebiasing of the varactor diodes 316 and 318 is conventional. The diode316 is connected in parallel with a capacitor 324 and the diode 318 isconnected in parallel with a capacitor 326. A node 319 connects thecathode of the varactor diode 316 to a resistor 320. A positive biasvoltage from a source not depicted in FIG. 3 is applied to the node 319via the resistor 320. A terminal 313 connected between the capacitors324 and 326 is also connected to the input of amplifier 10. The terminal313 is also connected between the varactor diodes 316 and 318. The anodeof the varactor diode 318 at terminal 332 is connected to ground via aresistor 334.

If the voltages across the varactor diodes 316 and 318 are respectivelyV_(A) and V_(B), if the capacitances of the varactor diodes 316 and 318are respectively C_(A) and C_(B), and if the equivalent capacitancedenoted by the value C of equation (A) given below is kept constant,then a variable capacitive voltage divider is created which will notappreciably pull the frequency of the oscillator 300 as the divide ratio{V_(A) /(V_(A) +V_(B))} is changed. This arrangement thus minimizes theinteractions between the ALC loop and the AFC loop (both to bedescribed).

    C=(C.sub.A * C.sub.B)/(C.sub.A +C.sub.B)                   (A)

The capacitance versus voltage characteristic of varactor diodes isconventionally classified by a parameter commonly denoted by y. If y isunity, equation A is automatically satisfied with the value of thecapacitors 324 and 326 set to zero. For higher values of y, such as 2,the values of the capacitors 324 and 326 can be chosen to allow equationA to be approximately obeyed. For example, with BB512 diodes, 150 pF forthe capacitors 324 and 326 will obey equation (A) within one percent(1%), where C_(A) is now the sum of the capacitances of the varactordiode 316 and the capacitor 324 and C_(B) is the sum of the capacitancesof the varactor diode 318 and the capacitor 326. The ability to usehigher y diodes is important because they are the only kind available inthe large values typically needed for the varactors diodes 316 and 318.

Depicted in FIG. 4 is another embodiment of the present inventionwherein an oscillator 400 includes a high precision enhanced ALCdetector. Oscillator 400 is similar to the oscillator 300 (shown in FIG.3) except the LC circuit 29 further includes a capacitor 402 connectedbetween node 50 and a terminal 404. The value of the inductor 28 isincreased so that the net inductive reactance of the series combinationof the inductor 28 and the capacitor 402 in the embodiment shown in FIG.4 is the same as the previous inductive reactance of the inductor 28 inFIG. 3. The voltage at the terminal 404 is stepped up compared to thevoltage at the node 50 in proportion to the loaded Q of the capacitor402 defined by the following equation (B):

    Q.sub.L =X.sub.C /(ESR/2)                                  (B)

where X_(C) is the reactance of the capacitor 402 and ESR is the ESR ofthe crystal resonator 30. The advantage of stepping up the voltage isthat the relative contribution of error due to thermal drift of thediode turn-on voltage is reduced because it is inversely proportional tosignal amplitude. The drift in turn-on voltage is the main source oftemperature induced error in diode detectors. If the capacitor 402 is azero-temperature coefficient type (which is commonly known as COG orNP0), the voltage across it for a given excitation current into thebridge circuit 31 will be constant in spite of temperature variations.So if the voltage at the terminal 404 is held invariant overtemperature, the current through the resonator 30 will also beinvariant. A potential disadvantage of this high precision ALC detectorin the oscillator 400 is that it substantially increases the uncorrectedfrequency drift of the oscillator because of the increased sensitivityof frequency to the inductance of the inductor 28 as its inductance israised. However, in the present invention, the oscillator is underclosed loop control of the bridge circuit 31 and the inductor 28 doesnot affect the balance frequency of the bridge circuit 31.

The input of the amplifier 38 is connected to the terminal 404. In theprior embodiment shown in FIG. 3, the input of the amplifier 38 isconnected directly to the bridge circuit 31 at the port 50. Theembodiment of FIG. 4 is preferred over that of FIG. 3 because itenhances several aspects of the AFC system. First, it provides a simpleway of implementing the 90 degree phase shift required to interfaceproperly with the synchronous demodulator 42. Second, it eliminates someproblems that occur in the previous embodiment when the bridge was farout of balance. A detailed description of the enhanced ALC detector willnow be presented below.

The reactance of the capacitor 402 should be chosen to be much largerthan the impedance at the port 50 of the bridge 31. A typical ratiowould be 20 to 1. Assuming this rule is followed, then the voltage atthe terminal 404 is given to a good approximation by the product of thecurrent through the capacitor 402 and the reactance of capacitor 402. Inother words, the magnitude of the voltage at the terminal 404 isproportional to the magnitude of the bridge current at terminal 50. Asdescribed previously, the ALC circuit takes advantage of this tostabilize bridge current. Furthermore, the phase of the voltage at theterminal 404 lags by almost exactly 90 degrees with respect to thebridge current, because of the capacitive reactance involved. This builtin 90 degree phase shift can be used instead introducing a 90 degreephase in the amplifiers 39, 40, and 41. The advantage of this, besidessimplicity, is that the 90 degree phase shift implemented by thecapacitor 402 is inherently more accurate than a phase shift implementedin the amplifiers 39, 40 and 41. When implemented as in FIG. 4, the AFCsynchronous detector 42 compares the phase of the bridge current withthe null port voltage at the terminal 52. In other words, the bridgecurrent provides the LO signal (after amplification by the amplifier38). This is in contrast to FIG. 3, where the bridge voltage at theterminal 50 is compared with the null port voltage at the terminal 52.

Using the bridge current instead of the bridge voltage as the LO signalsource eliminates two problems that occur when the bridge is far out ofbalance. The first problem is a side effect of the enhanced ALCdetector. When the bridge is far out of balance, the input impedance ofthe port 50 increases considerably above its balance value. With aconstant bridge current assured by the enhanced ALC detector, thevoltage at port 50 rises under severe out of balance conditions. If theLO signal were sourced from the terminal 50 (as in FIG. 3), itsamplitude would increase substantially when the bridge was out ofbalance. This considerably complicates the design of the synchronousdemodulator 42. By taking the signal from terminal 404, there isguaranteed to be no amplitude variation whatsoever because the voltageat this point is controlled by the ALC detector to be at a constantamplitude. A second problem is that the phase relationship between thebridge voltage and the null voltage shifts away from the plus or minus90 degree phase shift when near a null towards a 0 degree phase shift asthe bridge imbalance increases. This phase error results in improperoperation of the synchronous demodulator 42 such that the AFC detectormay not converge to resonance, when the terminal 50 is used as the LOsource. However, the bridge current always has a phase shift of plus orminus 90 degrees with respect to the null port voltage, no matter howfar the crystal is off resonance. Hence the enhanced AFC of FIG. 4always converges to resonance.

FIG. 5 depicts still another embodiment of the present invention wherethe transformer 34 shown in FIGS. 1-4 is eliminated and an electronicbridge 500 replaces the bridge circuit 31 shown in those prior figures.In FIG. 5, the crystal resonator 30 is connected to an input terminal502 of a precision phase inverter 504 (which will be described in moredetail). An output terminal 506 is connected to an image resistor 508.The image resistor 508 is connected to the resonator 30 and an invertinginput 510 of an operational amplifier 512. The inverting input 510 ofthe amplifier 512 is also connected to a resistor 516. The resistor 516is connected to an output terminal 518 of the operational amplifier 512.A non-inverting input 514 of the operational amplifier 512 is grounded.The electronic bridge 500 eliminates the magnetic susceptibility of themagnetic core of the transformer. The size of the transformer,particularly the height is typically greater than the components used inthe electronic bridge 500. The electronic bridge 500 is amenable tobeing implemented on a monolithic integrated circuit, which is a verydesirable attribute for commercial applications.

The theory of operation of the transformerless electronic bridge 500will now be presented. Like the previously described bridge, it mustmeet two requirements. The oscillator port must have an input impedancecharacteristic similar to a crystal and the null port must have an errorsignal such that the polarity of the phase when above resonance is theopposite of the polarity of the phase when below resonance. In FIG. 5,the operational amplifier 512 is configured as a transconductanceamplifier having an input impedance of essentially zero at its inputterminal 510. The voltage at the operational amplifier output terminal518 is equal to the input current divided by the resistance of thefeedback resistor 516. Since the input terminal 510 acts as a virtualground for the crystal 30, the input impedance of the oscillator port isequal to the shunt combination of the impedance of the crystal and theinput impedance of the precision phase inverter 504 (to be described ina later section). Since the impedance of the inverter 504 is very high,the input impedance of the oscillator port is practically the same asthe crystal impedance. Hence the first requirement for the bridge ismet.

With regard to the null port at the terminal 52, the situation atresonance is that the impedance of the crystal is a resistance equal tothe image resistor 508. The precision phase inverter 504 has an outputvoltage that is equal in magnitude and opposite in phase to its inputvoltage. Hence the voltage on the upper terminals of the crystal 30 andthe image resistor 508 are equal and opposite. Since the resistances areequal, the current flow through the crystal will be equal and oppositeto the current flow through the image resistor 508. Therefore, nocurrent will flow into the transconductance amplifier 512. The precisionphase inverter 504 effectively performs the function previously done bythe hybrid transformer. When the crystal is off resonance, the operationof the transformerless electronic bridge 500 is similar to thetransformer type bridge described previously, with the phase shiftbetween the ports being either 90 degrees ahead or behind depending onthe sign of the frequency error.

The accuracy and stability of the electronic bridge 500 depends largelyon the degree of precision of the phase inverter. In general, active RFphase inverters are far less precise than commonly available RF hybridtransformers. This is because the operating frequency (e.g., 10 MHz) isa significant percentage of the gain bandwidth product of even the mostwideband amplifiers currently available, as is well known in the fieldof active filters. To overcome this limitation, extraordinary means areneeded to stabilize the phase inverter. An active phase inverter canhave both amplitude and phase errors. Of the two, the phase error is themajor concern since phase is typically much more temperature sensitivethan amplitude.

Depicted in FIG. 6 is a detailed schematic of the precision phaseinverter 504 shown in FIG. 5. The phase inverter 504 uses bridgestabilization based on principles very similar to the previouslydescribed bridge stabilized oscillator. The difference is that thebridge is now simply a pair of matched resistors. Before a detaileddescription is given, an explanation will be provided about the theoryof operation of the precision phase inverter. The main signal path isfrom the input terminal 502 through an amplifier 604 with a nominalvoltage gain of preferably slightly more than +1, then through a secondamplifier 628 having a nominal voltage gain of preferably slightly morethan -1, and then to the output terminal 506. The first amplifier 604provides for a high input impedance as needed in the bridge 500 (in FIG.5), and the second amplifier 628 performs the basic phase inversion,again as needed by the bridge 500. An added feature of the firstamplifier 604 is that the phase can be adjusted by varying the biasvoltage across a varactor diode 616. Ideally, the input and outputvoltage should be equal and opposite. The input and output voltages 601& 603 are connected respectively to a matched pair of resistors 632 and638 of a half-bridge circuit 639 connected to an input terminal 633 ofan error amplifier 640. The input terminal 633 acts as a summing nodefor the currents through the matched resistors 632 and 638. If the inputand output voltages are equal and opposite, there will be no voltage atthe input terminal 633. If not, an error signal will be present having aphase with a polarity the same as the phase error between the signals atthe input 502 and the output 506. The phase of the error signal atterminal 642 is compared to the phase of the output signal at terminal634 by a special synchronous demodulator 636 and used to generate a DCvoltage that is integrated by an error integrator 645 and used to drivethe varactor diode 616. This feedback corrects the phase error bychanging the phase shift through the first amplifier 604. The varactordiode 616 works in conjunction with a feedback network comprising aresistor 612, a capacitor 610, a resistor 608, and the capacitor 614.

A detailed description of the precision phase inverter 504 will now begiven. The input terminal 502 is connected to non-inverting input 602 ofan operational amplifier 604 having a gain of about +1. An invertinginput 606 of the amplifier 604 is connected to a resistor 608, acompensation capacitor 610, another resistor 612, and the DC blockingcapacitor 614. The other end of the resistor 612 is grounded. The otherend of the capacitor 614 is connected to the cathode of a varactor diode616 and a resistor 618 which is used for biasing the varactor diode 616.The anode of the varactor diode 616 is grounded. An output terminal 620of the amplifier 604 is connected to the resistor 608, the capacitor 610and a resistor 622. An adjustable phase equalizer circuit 621 is formed.The other end of the resistor 622 is connected to a resistor 624 and aninverting input 626 of an operational amplifier 628 having a gain ofabout -1. A non-inverting input 627 of the amplifier 628 is grounded. Asa result, a non-precision inverter 629 is formed.

The output terminal 506 of the precision phase inverter 504 is connectedto an output of the amplifier 628, a reference amplifier 630, and oneend of the resistor 632. The amplifier 630 is connected to an LO input634 of a mixer 637. The other end of the resistor 632 is connected toone end of the resistor 638 and the amplifier 640. The amplifier 640 isconnected an RF input 642 of the mixer 637 which operates as asynchronous demodulator. An output 644 of the mixer 637 is connected toa resistor 646 whose other end is connected to a capacitor 648 and aninverting input 650 of an amplifier 652. A non-inverting input 654 ofthe amplifier 652 is grounded. An output 656 of the amplifier 652 isconnected to the other end of the biasing resistor 618. The inputterminal 502 of the precision phase inverter 504 is connected to theother end of the resistor 638.

In all of the preferred embodiments of the present invention, thecrystal resonator 30 is placed in an oven which operates to maintain thecrystal material at a prescribed temperature level. The oven enclosureand associated circuitry for heating, sensing, and controlling theinterior temperature of the oven can all be conventional and are notdepicted. An example of a suitable oven is the oven used for theaforementioned 10811 Oscillator made by HP, but modified to enclose thepresent invention. However, one advantage of the preferred embodimentsof the present invention is that only the crystal resonator is requiredto be in an oven and the remaining circuitry can be exposed to ambienttemperatures of the surroundings. In the 10811 Oscillator, the oven isalso used to enclose the other oscillator circuitry connected to theovenized crystal resonator in order to achieve the desired levels offrequency stability. In a working embodiment of the present invention,the frequency stability was comparable to the that of the 10811Oscillator but only the crystal resonator of the present invention wasovenized while the associated other circuitry was left exposed toambient temperatures.

FIGS. 7-11 depict schematics of an actual working embodiment of theoscillator circuit made in accordance with the teachings of the presentinvention. Specific component values are given. FIG, 7 shows a VCXOstage and FIG. 8 depicts a bridge circuit. FIG. 9 shows an ALC circuitand FIG. 10 depicts an AFC RF amplifier circuit. FIG. 11 shows asynchronous detector and an AFC DC amplifier circuit.

While the present invention has been described and illustrated withreference to specific embodiments, those skilled in the art willrecognize that modifications and variations may be made withoutdeparting from the principles disclosed by the teachings of the presentinvention. For example, the present invention can be used as part of animproved temperature compensated crystal oscillator (TCXO). As furtherexplanation of this alternate embodiment of the present invention, aseparate circuit senses the temperature of the ambient environment ofthe crystal. The temperature information is then used to compensate forany temperature induced drift occurring in the oscillator signalproduced by the present invention. A conventional TCXO is a well knowndevice that eliminates the need for an oven. Since an oven typicallyrequires a significant amount of power, the conventional and improvedTCXO's are useful for those applications where power is limited.

What is claimed is:
 1. An oscillator circuit for stabilizing anoscillator signal having a variable frequency comprising:a resonator foroperating at various vibration frequencies including a resonancefrequency and having a resonator impedance; a discriminating bridgecircuit being formed with the resonator as one arm and having first andsecond nodes; the bridge circuit responding to the resonator vibrationfrequencies to produce at the first node a first bridge signal with aphase and at the second node a second bridge signal with a phase shiftwith respect to the phase of the first bridge signal; the bridge circuitbeing so constructed and arranged that the phase shift leads or lags thephase of the first bridge signal by a first phase shift when theresonator is operating above or below its resonance frequency; thebridge circuit also having a bridge impedance at the first node, whereinthe bridge impedance has the same characteristics as the resonatorimpedance when the resonator is operating as a free-standing device;phase shifting circuitry for conditioning the second bridge signal sothat the phase shift is further shifted by a second phase shift andproducing an RF signal; a synchronous demodulator stage for respondingto both the first bridge and the RF signals and producing an errorsignal; the error signal having states respectively representing whenthe resonator is vibrating above, equal to and below its resonancefrequency; a control and oscillator stage having first, second and thirdterminals; the first terminal connected to receive a control signalderived from the error signal; the second terminal coupled to theresonator via the first node of the bridge circuit; the control andoscillator stage operating to produce the oscillator signal at the thirdterminal; the control and oscillator stage further being responsive tothe states of the error signal so that when the resonator is initiallyvibrating above or below its resonance frequency the vibration frequencyof the resonator will subsequently be reduced or increased respectivelytowards the resonance frequency; the vibration frequency of theresonator being unchanged when vibrating at its resonance frequency. 2.The oscillator circuit of claim 1 further comprising a secondsynchronous demodulator stage, an image resistor forming another arm ofthe bridge circuit, a voltage controlled (VC) resistance connected inseries with the image resistor resulting in a combined resistance, and aninety (90) degree phase shifting circuit coupled to receive the RFsignal for producing a second RF signal having a ninety degree phaseshift with respect to the RF signal; the second demodulator stageoperating in quadrature and being responsive to the first bridge signaland the second RF signal for producing a second demodulator signal forchanging the VC resistance so that the combined resistance of the VCresistance and the image resistor is equal to the ESR of the resonator.3. The oscillator circuit of claim 1 wherein the demodulator stagefurther comprises a mixer having first and second input mixer terminalsand an output mixer terminal, the first mixer input terminal beingcoupled to the first node, a fixed phase shifter having an outputconnected to the second input mixer terminal, and an integrator beingconnected to the output mixer terminal; the fixed phase shifter havingan input connected to the second node; the mixer operating for producinga DC signal at the output mixer terminal; and the integrator beingresponsive to the DC signal to produce the control signal.
 4. Theoscillator circuit of claim 1 wherein the control and oscillator stagefurther includes a resonant series LC circuit coupled between the firstand second terminals; the LC circuit having a variable reactance andbeing coupled to respond to the control signal for changing itsreactance so that the vibration frequency of the resonator is returnedto the resonance frequency whenever the resonator is not vibrating atits resonance frequency; the LC circuit maintaining its reactancewhenever the resonator is vibrating at its resonance frequency.
 5. Theoscillator circuit of claim 4 wherein the LC circuit comprises avaractor diode, which is voltage controlled for changing itscapacitance, and an inductor in series with the varactor diode.
 6. Theoscillator circuit of claim 4 further including an ALC (Automatic LevelControl) circuit coupled to the bridge circuit and the control andoscillator stage for controlling a drive signal which is produced by thecontrol and oscillator stage and which excites the resonator to vibrate.7. The oscillator circuit of claim 6 wherein the ALC includes acapacitive voltage divider comprising 2 varactor diodes connected inseries and having a combined series capacitance, each varactor diodebeing reverse biased and a tuning voltage applied for tuning eachvaractor diode in opposite directions with respect to one another sothat the combined series capacitance remains constant.
 8. The oscillatorcircuit of claim 7 further comprising an enhanced ALC where a capacitorhaving two ends is disposed such that one end is connected in serieswith an inductor which in turn is in series with a varactor diode; theother end of the capacitor being connected to the first node of thebridge so that the capacitor is located between the first node and thesecond terminal of the control and oscillator stage.
 9. The oscillatorcircuit of claim 6 further comprising an enhanced ALC where a capacitorhaving two ends is disposed such that one end is connected in serieswith an inductor which in turn is in series with a varactor diode; theother end of the capacitor being connected to the first node of thebridge so that the capacitor is located between the first node and thesecond terminal of the control and oscillator stage.
 10. The oscillatorcircuit of claim 1 further including an ALC (Automatic Level Control)circuit coupled to the bridge circuit and the control and oscillatorstage for controlling a drive signal which is produced by the controland oscillator stage and which excites the resonator to vibrate.
 11. Theoscillator circuit of claim 10 wherein the ALC includes a capacitivevoltage divider comprising 2 varactor diodes connected in series andhaving a combined series capacitance, each varactor diode being reversebiased and a tuning voltage applied for tuning each varactor diode inopposite directions with respect to one another so that the combinedseries capacitance remains constant.
 12. The oscillator circuit of claim11 further comprising an enhanced ALC where a capacitor having two endsis disposed such that one end is connected in series with an inductorwhich in turn is in series with a varactor diode; the other end of thecapacitor being connected to the first node of the bridge so that thecapacitor is located between the first node and the second terminal ofthe control and oscillator stage.
 13. The oscillator circuit of claim 10further comprising an enhanced ALC where a capacitor having two ends isdisposed such that one end is connected in series with an inductor whichin turn is in series with a varactor diode; the other end of thecapacitor being connected to the first node of the bridge so that thecapacitor is located between the first node and the second terminal ofthe control and oscillator stage.
 14. The oscillator circuit of claim 1wherein the bridge circuit is a half-lattice configuration comprising acenter-tapped transformer, an image resistor, and the resonator eachforming arms of the bridge circuit; wherein the resonator includes acrystal and an ESR (Equivalent Series Resistance); wherein the imageresistor has a resistance equal to the ESR; wherein the first and secondphase shifts are each ninety (90) degrees which results in the RF signalhaving a zero (0) phase shift with respect to the bridge signal when thevibration frequency of the resonator is below its resonance frequency,and which results in the RF signal having one hundred eighty (180)degrees phase shift with respect to the phase of the bridge signal whenthe vibration frequency of the resonator is above its resonancefrequency.
 15. The oscillator circuit of claim 14 further comprising asecond synchronous demodulator stage, a voltage controlled (VC)resistance connected in series with the image resistor resulting in acombined resistance, and a ninety (90) degree phase shifting circuitcoupled to receive the RF signal for producing a second RF signal havinga ninety degree phase shift with respect to the RF signal; the seconddemodulator stage operating in quadrature and being responsive to thefirst bridge signal and the second RF signal for producing a seconddemodulator signal for changing the VC resistance so that the combinedresistance of the VC resistance and the image resistor is equal to theESR of the resonator.
 16. The oscillator circuit of claim 14 furtherincluding an ALC (Automatic Level Control) circuit coupled to the bridgecircuit and the control and oscillator stage for controlling a drivesignal which is produced by the control and oscillator stage and whichexcites the resonator to vibrate.
 17. The oscillator circuit of claim 16wherein the ALC includes a capacitive voltage divider comprising 2varactor diodes connected in series and having a combined seriescapacitance, each varactor diode being reverse biased and a tuningvoltage applied for tuning each varactor diode in opposite directionswith respect to one another so that the combined series capacitanceremains constant.
 18. The oscillator circuit of claim 17 furthercomprising an enhanced ALC where a capacitor having two ends is disposedsuch that one end is connected in series with an inductor which in turnis in series with a varactor diode; the other end of the capacitor beingconnected to the first node of the bridge so that the capacitor islocated between the first node and the second terminal of the controland oscillator stage.
 19. The oscillator circuit of claim 16 furthercomprising an enhanced ALC where a capacitor having two ends is disposedsuch that one end is connected in series with an inductor which in turnis in series with a varactor diode; the other end of the capacitor beingconnected to the first node of the bridge so that the capacitor islocated between the first node and the second terminal of the controland oscillator stage.
 20. The oscillator circuit of claim 14 wherein thecontrol and oscillator stage further includes a resonant series LCcircuit coupled between the first and second terminals; the LC circuithaving a variable reactance and being coupled to respond to the controlsignal for changing its reactance so that the vibration frequency of theresonator is returned to the resonance frequency whenever the resonatoris not vibrating at its resonance frequency; the LC circuit maintainingits reactance whenever the resonator is vibrating at its resonancefrequency.
 21. The oscillator circuit of claim 20 further comprising asecond synchronous demodulator stage, a voltage controlled (VC)resistance connected in series with the image resistor resulting in acombined resistance, and a ninety (90) degree phase shifting circuitcoupled to receive the RF signal for producing a second RF signal havinga ninety degree phase shift with respect to the RF signal; the seconddemodulator stage operating in quadrature and being responsive to thefirst bridge signal and the second RF signal for producing a seconddemodulator signal for changing the VC resistance so that the combinedresistance of the VC resistance and the image resistor is equal to theESR of the resonator.
 22. The oscillator circuit of claim 20 wherein thedemodulator stage further comprises a mixer having first and secondinput mixer terminals and an output mixer terminal, the first mixerinput terminal being coupled to the first node, a fixed phase shifterhaving an output connected to the second input mixer terminal, and anintegrator being connected to the output mixer terminal; the fixed phaseshifter having an input connected to the second node; the mixeroperating for producing a DC signal at the output mixer terminal; andthe integrator being responsive the the DC signal to produce the controlsignal.
 23. The oscillator circuit of claim 20 wherein the LC circuitcomprises a varactor diode, which is voltage controlled for changing itscapacitance, and an inductor in series with the varactor diode.
 24. Theoscillator circuit of claim 20 further including an ALC (Automatic LevelControl) circuit coupled to the bridge circuit and the control andoscillator stage for controlling a drive signal which is produced by thecontrol and oscillator stage and which excites the resonator to vibrate.25. The oscillator circuit of claim 24 wherein the ALC includes acapacitive voltage divider comprising 2 varactor diodes connected inseries and having a combined series capacitance, each varactor diodebeing reverse biased and a tuning voltage applied for tuning eachvaractor diode in opposite directions with respect to one another sothat the combined series capacitance remains constant.
 26. Theoscillator circuit of claim 25 further comprising an enhanced ALC wherea capacitor having two ends is disposed such that one end is connectedin series with an inductor which in turn is in series with a varactordiode; the other end of the capacitor being connected to the first nodeof the bridge so that the capacitor is located between the first nodeand the second terminal of the control and oscillator stage.
 27. Theoscillator circuit of claim 24 further comprising an enhanced ALC wherea capacitor having two ends is disposed such that one end is connectedin series with an inductor which in turn is in series with a varactordiode; the other end of the capacitor being connected to the first nodeof the bridge so that the capacitor is located between the first nodeand the second terminal of the control and oscillator stage.
 28. Theoscillator circuit of claim 1 wherein one arm of the bridge circuitbetween the first and second nodes comprises a precision phase inverterconnected in series with an image resistor, and the second node isconnected to a transconductance amplifier circuit.
 29. The oscillatorcircuit of claim 28 further comprising a second synchronous demodulatorstage, a voltage controlled (VC) resistance connected in series with theimage resistor resulting in a combined resistance, and a ninety (90)degree phase shifting circuit coupled to receive the RF signal forproducing a second RF signal having a ninety degree phase shift withrespect to the RF signal; the second demodulator stage operating inquadrature and being responsive to the first bridge signal and thesecond RF signal for producing a second demodulator signal for changingthe VC resistance so that the combined resistance of the VC resistanceand the image resistor is equal to the ESR of the resonator.
 30. Theoscillator circuit of claim 28 further including an ALC (Automatic LevelControl) circuit coupled to the bridge circuit and the control andoscillator stage for controlling a drive signal which is produced by thecontrol and oscillator stage and which excites the resonator to vibrate.31. The oscillator circuit of claim 30 wherein the ALC includes acapacitive voltage divider comprising 2 varactor diodes connected inseries and having a combined series capacitance, each varactor diodebeing reverse biased and a tuning voltage applied for tuning eachvaractor diode in opposite directions with respect to one another sothat the combined series capacitance remains constant.
 32. Theoscillator circuit of claim 31 further comprising an enhanced ALC wherea capacitor having two ends is disposed such that one end is connectedin series with an inductor which in turn is in series with a varactordiode; the other end of the capacitor being connected to the first nodeof the bridge so that the capacitor is located between the first nodeand the second terminal of the control and oscillator stage.
 33. Theoscillator circuit of claim 30 further comprising an enhanced ALC wherea capacitor having two ends is disposed such that one end is connectedin series with an inductor which in turn is in series with a varactordiode; the other end of the capacitor being connected to the first nodeof the bridge so that the capacitor is located between the first nodeand the second terminal of the control and oscillator stage.
 34. Theoscillator circuit of claim 28 wherein the precision phase invertercomprises an input, an output, an adjustable phase equalizer connectedbetween the input and a non-precision inverter, the non-precisioninverter connected between the phase equalizer and the output, ahalf-bridge circuit connected between the input and a third synchronousdemodulator, the third synchronous demodulator connected between anerror integrator and the half-bridge circuit, the error integratorconnected between the third synchronous demodulator and the phaseequalizer, and the half bridge circuit also connected between the outputand the third synchronous demodulator.
 35. The oscillator circuit ofclaim 34 wherein the control and oscillator stage further includes aresonant series LC circuit coupled between the first and secondterminals; the LC circuit having a variable reactance and being coupledto respond to the control signal for changing its reactance so that thevibration frequency of the resonator is returned to the resonancefrequency whenever the resonator is not vibrating at its resonancefrequency; the LC circuit maintaining its reactance whenever theresonator is vibrating at its resonance frequency.
 36. The oscillatorcircuit of claim 35 wherein the LC circuit comprises a varactor diode,which is voltage controlled for changing its capacitance, and aninductor in series with the varactor diode.
 37. The oscillator circuitof claim 35 further including an ALC (Automatic Level Control) circuitcoupled to the bridge circuit and the control and oscillator stage forcontrolling a drive signal which is produced by the control andoscillator stage and which excites the resonator to vibrate.
 38. Theoscillator circuit of claim 37 wherein the ALC includes a capacitivevoltage divider comprising 2 varactor diodes connected in series andhaving a combined series capacitance, each varactor diode being reversebiased and a tuning voltage applied for tuning each varactor diode inopposite directions with respect to one another so that the combinedseries capacitance remains constant.
 39. The oscillator circuit of claim38 further comprising an enhanced ALC where a capacitor having two endsis disposed such that one end is connected in series with an inductorwhich in turn is in series with a varactor diode; the other end of thecapacitor being connected to the first node of the bridge so that thecapacitor is located between the first node and the second terminal ofthe control and oscillator stage.
 40. The oscillator circuit of claim 37further comprising an enhanced ALC where a capacitor having two ends isdisposed such that one end is connected in series with an inductor whichin turn is in series with a varactor diode; the other end of thecapacitor being connected to the first node of the bridge so that thecapacitor is located between the first node and the second terminal ofthe control and oscillator stage.
 41. The oscillator circuit of claim 28wherein the control and oscillator stage further includes a resonantseries LC circuit coupled between the first and second terminals; the LCcircuit having a variable reactance and being coupled to respond to thecontrol signal for changing its reactance so that the vibrationfrequency of the resonator is returned to the resonance frequencywhenever the resonator is not vibrating at its resonance frequency; theLC circuit maintaining its reactance whenever the resonator is vibratingat its resonance frequency.
 42. The oscillator circuit of claim 41wherein the LC circuit comprises a varactor diode, which is voltagecontrolled for changing its capacitance, and an inductor in series withthe varactor diode.
 43. The oscillator circuit of claim 41 furtherincluding an ALC (Automatic Level Control) circuit coupled to the bridgecircuit and the control and oscillator stage for controlling a drivesignal which is produced by the control and oscillator stage and whichexcites the resonator to vibrate.
 44. The oscillator circuit of claim 43wherein the ALC includes a capacitive voltage divider comprising 2varactor diodes connected in series and having a combined seriescapacitance, each varactor diode being reverse biased and a tuningvoltage applied for tuning each varactor diode in opposite directionswith respect to one another so that the combined series capacitanceremains constant.
 45. The oscillator circuit of claim 44 furthercomprising an enhanced ALC where a capacitor having two ends is disposedsuch that one end is connected in series with an inductor which in turnis in series with a varactor diode; the other end of the capacitor beingconnected to the first node of the bridge so that the capacitor islocated between the first node and the second terminal of the controland oscillator stage.
 46. The oscillator circuit of claim 43 furthercomprising an enhanced ALC where a capacitor having two ends is disposedsuch that one end is connected in series with an inductor which in turnis in series with a varactor diode; the other end of the capacitor beingconnected to the first node of the bridge so that the capacitor islocated between the first node and the second terminal of the controland oscillator stage.
 47. Method for stabilizing an oscillator signalgenerated from an oscillator circuit comprising the steps of:operating aresonator at variable vibration frequencies including a resonancefrequency; connecting the resonator to form one arm of a discriminatingbridge circuit, wherein the resonator when free standing outside thebridge circuit has impedance characteristics dependent upon itsvibration frequency; and the bridge circuit has an impedance; arrangingthe bridge circuit so that its impedance at a first node of the bridgecircuit has the same impedance characteristics as the free standingresonator; producing at the first node of the bridge circuit a firstbridge signal having a phase; producing at a second node of the bridgecircuit a second bridge signal having a phase shift which leads or lagsthe phase of the first bridge signal by a first prescribed phase shiftwhen the resonator is vibrating respectively above or below itsresonance frequency; phase shifting the second bridge signal by a secondprescribed phase shift and generating an RF signal; mixing the firstbridge and the RF signals in a synchronous demodulator stage andgenerating an error signal having states representing when the resonatoris vibrating above, equal to and below its resonance frequency;converting the error signal to a control signal and transmitting thecontrol signal to an oscillator stage having first, second, and thirdterminals; operating the oscillator stage to respond to the states ofthe control signal being applied to the first terminal so that when theresonator is initially vibrating above, equal to, or below its resonancefrequency the vibrating frequency of the resonator will subsequently bereduced, kept unchanged, or increased respectively towards the resonancefrequency; connecting the second terminal to the first node; andoperating the oscillator stage to respond to the resonator forgenerating the oscillator signal at the third terminal.
 48. The methodof claim 47 including the steps of providing an image resistor asanother arm of the bridge circuit; connecting a voltage controlled (VC)resistance in series with the image resistor and resulting in a combinedresistance; connecting a ninety (90) degree phase shifting circuit toreceive the RF signal for generating a second RF signal having a ninety(90) degree phase shift with respect to the RF signal; transmitting thefirst bridge signal and the second RF signal to a second synchronousdemodulator stage; operating the second synchronous demodulator stage inquadrature for producing a second demodulator signal; and responding tothe second demodulator signal and changing the VC resistance so that thecombined resistance is equal to the ESR of the resonator.
 49. The methodof claim 47 further comprising the steps of including in the demodulatora mixer having first and second input mixer terminals and an outputmixer terminal; connecting the first node to the first input mixerterminal; connecting an output of a fixed phase shifter to the secondinput mixer terminal; connecting an integrator to the output mixerterminal; connecting an input of the fixed phase shifter to the secondnode; operating the mixer for producing a DC signal at the output mixerterminal; and operating the integrator to respond to the DC signal forgenerating the control signal.
 50. The method of claim 47 furthercomprising the steps of including in the oscillator stage a resonantseries LC circuit having a variable reactance; coupling the LC circuitbetween the first and second terminals; responding to the control signalby changing the variable reactance so that the vibration frequency ofthe resonator is returned to the resonance frequency whenever theresonator is not vibrating at its resonance frequency; and maintainingthe reactance unchanged whenever the resonator is vibrating at itsresonance frequency.
 51. The method of claim 50 further comprising thesteps of including in the LC circuit a varactor diode having a voltagecontrolled capacitance and an inductor; connecting the diode andinductor in series; and operating the diode for responding to thecontrol signal.
 52. The method of claim 51 including the steps ofconnecting an ALC (Automatic Level Control) circuit to the bridgecircuit and the oscillator stage for controlling a drive signal producedby the oscillator stage; and transmitting the drive signal to theresonator for exciting it to vibrate.
 53. The method of claim 52including the step of connecting a capacitor between the second terminaland the first node so as to produce an enhanced ALC circuit.
 54. Themethod of claim 47 including the steps of connecting an ALC (AutomaticLevel Control) circuit to the bridge circuit and the oscillator stagefor controlling a drive signal produced by the oscillator stage; andtransmitting the drive signal to the resonator for exciting it tovibrate.
 55. The method of claim 54 including the steps of connectingone end of a capacitor to on end of an inductor; connecting the otherend of the inductor in series with a varactor diode; connecting theother end of the capacitor to the first node so that capacitor islocated between the first node and the second terminal of the oscillatorstage.
 56. The method of claim 54 including the steps of providing acapacitive voltage divider in the ALC circuit comprising two varactordiodes connected in series and having a combined capacitance; reversebiasing each diode; tuning each diode in opposite directions withrespect to one another with a tuning voltage so that the combined seriescapacitance remains constant.
 57. The method of claim 47 including thesteps of connecting between the first and second nodes a precision phaseinverter in series with an image resistor; and connecting atransconductance amplifier circuit to the second node.
 58. The method ofclaim 57 further comprising the steps of connecting an adjustable phaseequalizer between an input of the precision phase inverter and anon-precision inverter; connecting the non-precision inverter betweenthe phase equalizer and an output of the precision phase inverter;connecting a half-bridge circuit between the input and a thirdsynchronous demodulator; connecting the third synchronous demodulatorbetween an error integrator and the half-bridge circuit; connecting theerror integrator between the third synchronous demodulator and the phaseequalizer; and connecting the half-bridge circuit between the output andthe third synchronous demodulator.
 59. The method of claim 58 furthercomprising the steps of including in the oscillator stage a resonantseries LC circuit having a variable reactance; coupling the LC circuitbetween the first and second terminals; responding to the control signalby changing the variable reactance so that the vibration frequency ofthe resonator is returned to the resonance frequency whenever theresonator is not vibrating at its resonance frequency; and maintainingthe reactance unchanged whenever the resonator is vibrating at itsresonance frequency.
 60. The method of claim 47 further comprising thesteps of configuring the bridge circuit as a half-lattice wherein acenter-tapped transformer, an image resistor, and the resonator eachform arms of the bridge circuit; configuring the resonator as a crystalhaving and ESR (Equivalent Series Resistance); providing the imageresistor with a resistance equal to the ESR; making the first and secondprescribed phase shifts to be each equal to ninety (90) degreesresulting in the RF signal having a zero (0) phase shift with respect tothe bridge signal when the vibration frequency of the resonator is belowits resonance frequency, and which results in the RF signal having onehundred eighty (180) degrees phase shift with respect to the phase ofthe bridge signal when the vibration frequency of the resonator is aboveits resonance frequency.
 61. The method of claim 60 including the stepsof connecting a voltage controlled (VC) resistance in series with theimage resistor and resulting in a combined resistance; connecting aninety (90) degree phase shifting circuit to receive the RF signal forgenerating a second RF signal having a ninety (90) degree phase shiftwith respect to the RF signal; transmitting the first bridge signal andthe second RF signal to a second synchronous demodulator stage;operating the second synchronous demodulator stage in quadrature forproducing a second demodulator signal; and responding to the seconddemodulator signal and changing the VC resistance so that the combinedresistance is equal to the ESR of the resonator.
 62. The method of claim61 further comprising the steps of including in the oscillator stage aresonant series LC circuit having a variable reactance; coupling the LCcircuit between the first and second terminals; responding to thecontrol signal by changing the variable reactance so that the vibrationfrequency of the resonator is returned to the resonance frequencywhenever the resonator is not vibrating at its resonance frequency; andmaintaining the reactance unchanged whenever the resonator is vibratingat its resonance frequency.
 63. The method of claim 62 including thesteps configuring the LC circuit as varactor diode, which is voltagecontrolled for changing its capacitance; connecting an inductor inseries with the varactor diode.
 64. The method of claim 62 including thesteps of connecting an ALC (Automatic Level Control) circuit to thebridge circuit and the oscillator stage for controlling a drive signalproduced by the oscillator stage; and transmitting the drive signal tothe resonator for exciting it to vibrate.
 65. The method of claim 64including the steps of connecting a capacitor between the secondterminal and the first node so as to produce an enhanced ALC circuit.66. The method of claim 64 including the steps of providing a capacitivevoltage divider in the ALC circuit comprising two varactor diodesconnected in series and having a combined capacitance; reverse biasingeach diode; tuning each diode in opposite directions with respect to oneanother with a tuning voltage so that the combined series capacitanceremains constant.
 67. An oscillator comprising:a resonator having anEquivalent Series Resistance (ESR) and a resonant frequency that is afunction of an excitation current; a circuit for generating aprogrammable negative resistance; a resonant series LC circuit havinginductive and capacitive elements and a voltage appearing at a terminallocated between the inductive and capacitive elements; a voltagedetector connected to the terminal and used to program the value of thenegative resistance to be equal to the ESR, such that the negativeresistance produces a fixed value of the voltage at the terminalresulting in a desired value of current for exciting the resonator. 68.The oscillator of claim 67 wherein the circuit for generating furtherincludes a gain control stage comprising two varactor diodes connectedin series; wherein the two diodes have a combined series capacitancewith each diode being reverse biased and a tuning voltage is applied fortuning each diode in opposite directions with one another so that thecombined series capacitance remains constant.
 69. The oscillator ofclaim 68 wherein the voltage detector includes a diode.
 70. Theoscillator of claim 69 wherein the resonator includes a crystal. 71.Method for stabilizing an oscillator signal produced by an oscillatorcircuit comprising the steps of:exciting a resonator into resonantfrequency via the application of an excitation current; the resonatorhaving an Equivalent Series Resistance (ESR); generating a programmablenegative resistance; configuring a resonant series LC circuit withinductive and capacitive elements and generating a voltage at a terminallocated between the inductive and capacitive elements; responding to thevoltage for programming the value of the negative resistance to be equalto the ESR such that the negative resistance produces a fixed value ofthe voltage at the terminal resulting in a desired value of current forexciting the resonator.
 72. The method of claim 71 wherein the step ofgenerating a resistance includes connecting two varactor diodes inseries with a combined series capacitance; reverse biasing each diodewith a tuning voltage applied to each diode in opposite directions withone another so that the combined series capacitance remains constant.73. The method of claim 72 including the step of using a diode in thestep of responding.
 74. The method of claim 73 including the step ofconfiguring the resonator to include a crystal.